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LT3474 Step-Down 1A LED Driver
FEATURES

DESCRIPTIO
True Color PWM Delivers Constant Color with 400:1 Dimming Range Wide Input Range: 4V to 36V Up to 1A LED Current Adjustable 200kHz-2MHz Switching Frequency Adjustable Control of LED Current Integrated Boost Diode High Output Current Accuracy is Maintained Over a Wide Range from 35mA to 1A Open LED and Short-Circuit Protection High Side Sense Allows Grounded Cathode Connection Uses Small Inductors and Ceramic Capacitors Compact 16-Lead TSSOP Thermally Enhanced Surface Mount Package
The LT(R)3474 is a fixed frequency step-down DC/DC converter designed to operate as a constant-current source. An internal sense resistor monitors the output current allowing accurate current regulation, ideal for driving high current LEDs. High output current accuracy is maintained over a wide current range, from 35mA to 1A, allowing a wide dimming range. Unique PWM circuitry allows a dimming range of 400:1, avoiding the color shift normally associated with LED current dimming. The high switching frequency offers several advantages, permitting the use of small inductors and ceramic capacitors. Small inductors combined with the LT3474's 16 lead TSSOP surface mount package save space and cost versus alternative solutions. The constant switching frequency combined with low-impedance ceramic capacitors result in low, predictable output ripple. With its wide input range of 4V to 36V, the LT3474 regulates a broad array of power sources, from 4-cell batteries and 5V logic rails to unregulated wall transformers, lead acid batteries and distributed power supplies. A current mode PWM architecture provides fast transient response and cycle-by-cycle current limiting. Frequency foldback and thermal shutdown provide additional protection.
, LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patent Pending
APPLICATIO S

Automotive and Avionic Lighting Architectural Detail Lighting Display Backlighting Constant Current Sources
TYPICAL APPLICATIO
VIN 5V TO 36V
2.2F
Step-Down 1A LED Driver
95
VIN
VIN = 12V
BOOST
0.22F 10H
90 85 EFFICIENCY (%) 80
SHDN SW LT3474 RT REF
80.6k
0.1F BIAS OUT PWM LED DIMMING CONTROL 2.2F
75 70 65
VADJ VC GND
LED1
60 55 0 200 400 800 600 LED CURRENT (mA) 1000
3474 G02
3474 TA01a
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Efficiency
TWO SERIES CONNECTED WHITE LUXEON III STARS ONE WHITE LUXEON III STAR
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LT3474
ABSOLUTE
(Note 1)
AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW DNC* OUT LED VIN SW BOOST BIAS GND 1 2 3 4 5 6 7 8 17 16 DNC* 15 GND 14 PWM 13 VADJ 12 VC 11 REF 10 SHDN 9 RT
VIN Pin ....................................................... (-0.3V), 36V BIAS Pin .................................................................. 25V BOOST Pin Voltage .................................................. 51V BOOST above SW Pin .............................................. 25V OUT, LED Pins ......................................................... 15V PWM Pin ................................................................. 10V VADJ Pin ..................................................................... 6V VC, REF, RT Pins ........................................................ 3V SHDN Pin ..................................................................VIN BIAS Pin Current ....................................................... 1A Maximum Junction Temperature (Note 2) ............ 125C Operating Temperature Range (Note 3) ... -40C to 85C Storage Temperature Range .................. -65C to 150C Lead Temperature (Soldering, 10 sec).................. 300C
ORDER PART NUMBER LT3474EFE LT3474IFE
FE PACKAGE 16-LEAD PLASTIC TSSOP JC = 8C/W, JA = 40C/W EXPOSED PAD (PIN 17) IS GROUND AND MUST BE ELECTRICALLY CONNECTED TO THE PCB *DO NOT CONNECT EXTERNAL CIRCUITRY TO THESE PINS.
FE PART MARKING 3474EFE 3474IFE
Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges.
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3).
PARAMETER Minimum Input Voltage Input Quiescent Current Shutdown Current LED Pin Current Not Switching SHDN = 0.3V VADJ Tied to VREF
ELECTRICAL CHARACTERISTICS
CONDITIONS
MIN
TYP 3.5 2.6 0.01
MAX 4 4 2 1.02 1.025 0.207 0.210 1.265
UNITS V mA A A A A A V %/V %/A
VADJ Tied to VREF/5
0.98 0.968 0.193 0.186 1.230
1 0.2 1.25 0.01 0.0002
REF Voltage Reference Voltage Line Regulation Reference Voltage Load Regulation VADJ Pin Bias Current Switching Frequency Maximum Duty Cycle RT = 80.6k 5V < VIN < 36V 0 < IREF < 250A

20 470 450 90 500 95 76 98 70 2.60 2.65 10.3 0.9 0.8 100 100
400 530 540
RT = 80.6k RT = 10k RT = 232k RT = 80.6k, VOUT = 0V VSHDN = SHDN Threshold
Foldback Frequency SHDN Threshold (to Switch) SHDN Pin Current PWM Threshold VC Switching Threshold VC Source Current VC Sink Current
2.70 12.3 1.2
8.3 0.4
VC = 1V VC = 1V
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nA kHz kHz % % % kHz V A V V A A
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LT3474
ELECTRICAL CHARACTERISTICS
PARAMETER LED to VC Current Gain LED to VC Transresistance VC to Switch Current Gain VC Clamp Voltage VC Pin Current in PWM Mode OUT Pin Clamp Voltage OUT Pin Current in PWM Mode Switch Current Limit (Note 4) Switch VCESAT Boost Pin Current Switch Leakage Current Minimum Boost Voltage (Note 5) Boost Diode Forward Voltage IDIO = 100mA Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 3: The LT3474E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3474I is guaranteed to meet performance specifications over the -40C to 85C operating temperature range. ISW = 1A ISW = 1A VOUT = 4V, VPWM = 0.3V

The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3).
CONDITIONS MIN TYP 1.5 1 2 1.9 VC = 1V, VPWM = 0.3V
MAX
UNITS A/mA V/mA A/V V
0.01 13.2 1.6 13.8 0.1 2.1 380 30 0.01 1.9 600
1 14.5 10 3 500 50 1 2.5
A V A A mV mA A V mV
Note 4: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. Note 5: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch.
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LT3474 TYPICAL PERFOR A CE CHARACTERISTICS
LED Current vs VADJ
1000 TA = 25C 800
SWITCH VOLTAGE DROP (mV)
LED CURRENT (mA)
LED CURRENT (mA)
600
400
200
0
0
0.25
0.75 0.5 VADJ (V)
Current Limit vs Duty Cycle
2.5
2.5
2
CURRENT LIMIT (A)
CURRENT LIMIT (A)
1.5 MINIMUM 1
1.5
CURRENT LIMIT (A)
TYPICAL
0.5
0
0
20
60 40 DUTY CYCLE (%)
Oscillator Frequency vs RT
600
TA = 25C
OSCILLATOR FREQUENCY (kHz)
OSCILLATOR FREQUENCY (kHz)
550
1000
OSCILLATOR FREQUENCY (kHz)
100 10 RT (k)
100
3474 G09
4M
UW
1 80
3474 G06
LED Current vs Temperature
1200 VADJ = VREF 1000 800 600 400 200 0 -50 -25 VADJ = VREF/5
Switch Voltage Drop
700 TA = 25C 600 500 400 300 200 100 0
1.25
3474 GO3
50 25 75 0 TEMPERATURE (C)
100
125
0
1000 500 SWITCH CURRENT (mA)
1500
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Switch Current Limit vs Temperature
2.5
Current Limit vs Output Voltage
TA = 25C 2
2
1.5
1
1
0.5
0.5
100
0 -50
0
-25 50 25 0 75 TEMPERATURE (C) 100 125
0
2
4
6 VOUT (V)
8
10
12
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Oscillator Frequency vs Temperature
RT = 80.6k
600 500 400 300 200 100 0
Oscillator Frequency Foldback
TA = 25C RT = 80.6k
500
450
400 -50
-25
75 0 25 50 TEMPERATURE (C)
100
125
0
0.5
1
1.5 VOUT (V)
2
2.5
3474 G11
3474 G10
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LT3474 TYPICAL PERFOR A CE CHARACTERISTICS
Boost Pin Current
60 TA = 25C 50 BOOST PIN CURRENT (mA) INPUT CURRENT (mA) 40 30 20 10 0 2.5 2.0 1.5 1.0 0.5 0 0 250 500 750 1000 1250 SWITCH CURRENT (mA) 1500
1.240
VREF (V)
Schottky Reverse Leakage
20 VR = 5V FORWARD CURRENT (mA)
400 500
REVERSE CURRENT (A)
300
10
8
INPUT CURRENT
4
200
5
4
2
100
0 -50
0
-25
75 0 25 50 TEMPERATURE (C)
100
125
0
200 600 800 400 FORWARD VOLTAGE (mV)
100
3474 G19
0 0 6 12 18 VIN (V) 24 30
0 36
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Minimum Input Voltage, One White Luxeon III Star
6 5 4 TO RUN TA = 25C TO START
10
Minimum Input Voltage, Two Series Connected White Luxeon III Stars
TA = 25C 9
VIN (V)
VIN (V)
LED VOLTAGE 3 2 1 0 0 200 400 600 800 LED CURRENT (mA) 1000
3474 G17
8
TO START
7 TO RUN LED VOLTAGE 6
5
0
200
600 800 400 LED CURRENT (mA)
1000
3474 G18
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INPUT CURRENT (mA)
OUTPUT VOLTAGE (V)
15
UW
3473 G12
Quiescent Current
3.0 TA = 25C
1.255 1.260
Reference Voltage
1.250
1.245
0
6
12
18 VIN (V)
24
30
36
3474 G13
1.235 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
3474 G14
Schottky Forward Voltage Drop
TA = 25C
Open-Circuit Output Voltage and Input Current
16 TA = 25C 12 OUTPUT VOLTAGE 6 8
LT3474
PI FU CTIO S
DNC (Pins 1, 16): Do not connect external circuitry to these pins. OUT (Pin 2): The OUT pin is the input to the current sense resistor. Connect this pin to the inductor and the output capacitor. LED (Pin 3): The LED pin is the output of the current sense resistor. Connect the anode of the LED here. VIN (Pin 4): The VIN pin supplies current to the LT3474's internal circuitry and to the internal power switch and must be locally bypassed. SW (Pin 5): The SW pin is the output of the internal power switch. Connect this pin to the inductor and switching diode. BOOST (Pin 6): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. BIAS (Pin 7): The BIAS pin connects through a Schottky diode to BOOST. Tie to OUT. GND (Pins 8, 15, Exposed Pad Pin 17): Ground. Tie both GND pins and the Exposed Pad directly to the ground plane. The Exposed Pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. It must be soldered to the circuit board for proper operation. RT (Pin 9): The RT pin is used to set the internal oscillator frequency. Tie an 80.6k resistor from RT to GND for a 500kHz switching frequency. SHDN (Pin 10): The SHDN pin is used to shut down the switching regulator and the internal bias circuits. The 2.6V switching threshold can function as an accurate undervoltage lockout. Pull below 0.3V to shut down the LT3474. Pull above 2.65V to enable the LT3474. Tie to VIN if the SHDN function is unused. REF (Pin 11): The REF pin is the buffered output of the internal reference. Either tie the REF pin to the VADJ pin for a 1A output current, or use a resistor divider to generate a lower voltage at the VADJ pin. Leave this pin unconnected if unused. VC (Pin 12): The Vc pin is the output of the internal error amp. The voltage on this pin controls the peak switch current. Use this pin to compensate the control loop. VADJ (Pin 13): The VADJ pin is the input to the internal voltage to current amplifier. Connect the VADJ pin to the REF pin for a 1A output current. For lower output currents, program the VADJ pin using the following formula: ILED = 1A * VADJ/1.25V. PWM (Pin 14): The PWM pin controls the connection of the VC pin to the internal circuitry. When the PWM pin is low, the VC pin is disconnected from the internal circuitry and draws minimal current. If the PWM feature is unused, leave this pin unconnected.
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LT3474
BLOCK DIAGRA W
BIAS SHDN INT REG AND UVLO 7 SLOPE COMP RT
VIN CIN
4
VIN
10
R C1 S Q Q DRIVER
BOOST
6 C1
Q1 SW 5 D1
L1
9 RT
OSC
FREQUENCY FOLDBACK OUT 2 C2 100 0.1 LED 2V 3 DLED1
11
REF
14
PWM VADJ Q2
13
+
1.25V gm VC 1.25k GND 8
-
12 CC
3474 BD
Figure 1. Block Diagram
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LT3474
APPLICATIO S I FOR ATIO
Operation
The LT3474 is a constant frequency, current mode regulator with an internal power switch capable of generating a constant 1A output. Operation can be best understood by referring to the Block Diagram. If the SHDN pin is tied to ground, the LT3474 is shut down and draws minimal current from the input source tied to VIN. If the SHDN pin exceeds 1.5V, the internal bias circuits turn on, including the internal regulator, reference, and oscillator. The switching regulator will only begin to operate when the SHDN pin exceeds 2.65V. The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-by-cycle current limit. A pulse from the oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC, current comparator C1 resets the flip-flop, turning off the switch. The current in the inductor flows through the external Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output current by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.9V, and an active clamp of 1.9V limits the output current. The voltage on the VADJ pin sets the current through the LED pin. The NPN Q2 pulls a current proportional to the voltage on the VADJ pin through the 100 resistor. The gm amplifier servos the VC pin to set the current through the 0.1 resistor and the LED pin. When the voltage drop
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across the 0.1 resistor is equal to the voltage drop across the 100 resistor, the servo loop is balanced. Tying the REF pin to the VADJ pin sets the LED pin current to 1A. Tying a resistor divider to the REF pin allows the programming of LED pin currents of less than 1A. LED pin current can also be programmed by tying the VADJ pin directly to a voltage source. An LED can be dimmed with pulse width modulation using the PWM pin and an external NFET. If the PWM pin is unconnected or pulled high, the part operates nominally. If the PWM pin is pulled low, the VC pin is disconnected from the internal circuitry and draws minimal current from the compensation capacitor. Circuitry drawing current from the OUT pin is also disabled. This way, the VC pin and the output capacitor store the state of the LED pin current until PWM is pulled high again. This leads to a highly linear relationship between pulse width and output light, allowing for a large and accurate dimming range. The RT pin allows programming of the switching frequency. For applications requiring the smallest external components possible, a fast switching frequency can be used. If very low or very high input voltages are required, a slower switching frequency can be programmed. During startup VOUT will be at a low voltage. The NPN Q2 can only operate correctly with sufficient voltage at VOUT, around 1.7V. A comparator senses VOUT and forces the VC pin high until VOUT rises above 2V, and Q2 is operating correctly. The switching regulator performs frequency foldback during overload conditions. An amplifier senses when VOUT is less than 2V and begins decreasing the oscillator frequency down from full frequency to 20% of the nominal frequency when VOUT = 0V. The OUT pin is less than 2V during startup, short circuit, and overload conditions. Frequency foldback helps limit switch current under these conditions.
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LT3474
APPLICATIO S I FOR ATIO
The switch driver operates either from VIN or from the BOOST pin. An external capacitor and internal Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. Open Circuit Protection The LT3474 has internal open circuit protection. If the LED is absent or fails open, the LT3474 clamps the voltage on the LED pin at 14V. The switching regulator then skips cycles to limit the input current. The output voltage during an open LED condition is shown in the typical performance section. Undervoltage Lockout Undervoltage lockout (UVLO) is typically used in situations where the input supply is current limited, or has high source resistance. A switching regulator draws constant power from the source, so the source current increases as the source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. An internal comparator will force the part into shutdown when VIN falls below 3.5V. If an adjustable UVLO threshold
VIN R1 SHDN VIN 2.65V
C1
R2
Figure 2. Undervoltage Lockout
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is required, the SHDN pin can be used. The threshold voltage of the SHDN pin comparator is 2.65V. A internal resistor pulls 10.3A to ground from the SHDN pin at the UVLO threshold. Choose resistors according to the following formula:
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R2 =
2.65V VTH - 2.65V - 10.3A R1
VTH = UVLO Threshold Example: Switching should not start until the input is above 8V. VTH = 8V R1 = 100k
R2 = 2.65V = 61.9k 8 V - 2.65V - 10.3A 100k
Keep the connections from the resistors to the SHDN pin short and make sure the coupling to the SW and BOOST pins is minimized. If high resistance values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from switching nodes.
LT3474
VC
10.3A GND
3474 F02
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LT3474
APPLICATIO S I FOR ATIO
Setting the Switching Frequency The LT3474 uses a constant frequency architecture that can be programmed over a 200kHz to 2MHz range with a single external timing resistor from the RT pin to ground. The current that flows into the timing resistor is used to charge an internal oscillator capacitor. A graph for selecting the value of RT for a given operating frequency is shown in the Typical Performance Characteristics section. Table 1 shows suggested RT selections for a variety of switching frequencies.
Table 1. Switching Frequencies
SWITCHING FREQUENCY (MHz) 2 1.5 1 0.7 0.5 0.3 0.2 RT (k) 10 18.7 33.2 52.3 80.6 147 232
Operating Frequency Selection The choice of operating frequency is determined by several factors. There is a tradeoff between efficiency and component size. Higher switching frequency allows the use of smaller inductors at the cost of increased switching losses and decreased efficiency. Another consideration is the maximum duty cycle. In certain applications, the converter needs to operate at a high duty cycle in order to work at the lowest input voltage possible. The LT3474 has a fixed oscillator off-time and a variable on-time. As a result, the maximum duty cycle increases as the switching frequency is decreased. Input Voltage Range The minimum operating voltage is determined either by the LT3474's undervoltage lockout of 4V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages:
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DC =
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( VIN - VSW + VF )
( VOUT + VF )
where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.3V at maximum load). This leads to a minimum input voltage of:
VIN(MIN) = VOUT + VF - VF + VSW DCMAX
with DCMAX = 1-tOFF(MIN) * f where t0FF(MIN) is equal to 200ns and f is the switching frequency. Example: f = 500kHz, VOUT = 4V
DCMAX = 1- 200ns * 500kHz = 0.90 4V + 0.4V - 0.4V + 0.4V = 4.9 V VIN(MIN) = 9 0.9
The maximum operating voltage is determined by the absolute maximum ratings of the VIN and BOOST pins, and by the minimum duty cycle.
VIN(MAX ) =
VOUT + VF - VF + VSW DCMIN
with DCMIN = tON(MIN) * f where tON(MIN) is equal to 160ns and f is the switching frequency. Example: f = 500kHz, VOUT = 2.5V
DCMIN = 160ns * 500kHz = 0.08 2.5V + 0.4V VIN(MAX ) = - 0.4V + 0.4V = 36 V 0.08 0
The minimum duty cycle depends on the switching frequency. Running at a lower switching frequency might allow a higher maximum operating voltage. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the Absolute Maximum Rating.
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LT3474
APPLICATIO S I FOR ATIO
Inductor Selection and Maximum Output Current A good first choice for the inductor value is L = ( VOUT + VF ) * 900kHz f
where VF is the voltage drop of the catch diode (~0.4V), f is the switching frequency and L is in H. With this value the maximum load current will be 1.1A, independent of input voltage. The inductor's RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.2. Table 2 lists several vendors and types that are suitable. For robust operation at full load and high input voltages (VIN > 30V), use an inductor with a saturation current higher than 2A.
Table 2. Inductors
PART NUMBER Sumida CR43-3R3 CR43-4R7 CDRH4D16-3R3 CDRH4D28-3R3 CDRH4D28-4R7 CDRH5D28-100 CDRH5D28-150 CDRH73-100 CDRH73-150 Coilcraft DO1606T-332 DO1606T-472 DO1608C-332 DO1608C-472 MOS6020-332 MOS6020-472 3.3 4.7 3.3 4.7 3.3 10 1.30 1.10 2.00 1.50 1.80 1.50 0.100 0.120 0.080 0.090 0.046 0.050 2.0 2.0 2.9 2.9 2.0 2.0 3.3 4.7 3.3 3.3 4.7 10 15 10 15 1.44 1.15 1.10 1.57 1.32 1.30 1.10 1.68 1.33 0.086 0.109 0.063 0.049 0.072 0.048 0.076 0.072 0.130 3.5 3.5 1.8 3.0 3.0 3.0 3.0 3.4 3.4 VALUE (H) IRMS (A) DCR () HEIGHT (mm)
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The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology's Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid sub-harmonic oscillations. See Application Note 19. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-topeak inductor ripple current. The LT3474 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3474 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor IL =
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(1- DC)( VOUT + VF ) (L * f)
where f is the switching frequency of the LT3474 and L is the value of the inductor. The peak inductor and switch current is ISW(PK ) = IL(PK ) = IOUT + IL 2
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LT3474
APPLICATIO S I FOR ATIO
To maintain output regulation, this peak current must be less than the LT3474's switch current limit ILIM. For SW1, ILIM is at least 1.6A at low duty cycles and decreases linearly to 1.15A at DC = 0.8. The maximum output current is a function of the chosen inductor value:
IOUT(MAX ) = ILIM -
IL 2
= 1.6 A * 1- 0.35 * DC -
(
)
IL 2
Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3474 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than IL/2. Input Capacitor Selection Bypass the input of the LT3474 circuit with a 2.2F or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
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capacitor is required to reduce the resulting voltage ripple at the LT3474 input and to force this switching current into a tight local loop, minnimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. The RMS input is:
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CINRMS = IOUT *
VOUT VIN - VOUT VIN
(
) < IOUT
2
and is largest when VIN = 2VOUT (50% duty cycle). Considering that the maximum load current is 1A, RMS ripple current will always be less that 0.5A The high switching frequency of the LT3474 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 10F. The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types. An alternative to a high value ceramic capacitor is a lower value ceramic along with a larger electrolytic capacitor. The electrolytic capacitor likely needs to be greater than 10F in order to meet the ESR and ripple current requirements. The input capacitor is likely to see high surge currents when the input source is applied. Tantalum capacitors can fail due to an over-surge of current. Only use tantalum capacitors with the appropriate surge current rating. The manufacturer may also recommend operation below the rated voltage of the capacitor.
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LT3474
APPLICATIO S I FOR ATIO
A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source), this tank can ring, doubling the input voltage and damaging the LT3474. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Applicaion Note 88. Output Capacitor Selection For most LEDs, a 2.2F 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. Other types and values will also work; the following discusses tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilizes the LT3474's control loop. Because the LT3474 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option.
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You can estimate output ripple with the following equation:
VRIPPLE = IL (8 * f * COUT ) for ceramic capacitors
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where IL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula:
IC(RMS) =
IL 12
The low ESR and small size of ceramic capacitors make them the preferred type for LT3474 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Table 3 lists several capacitor vendors.
Table 3. Low-ESR Surface Mount Capacitors
VENDOR Taiyo-Yuden AVX TDK TYPE Ceramic Ceramic Ceramic SERIES X5R, X7R X5R, X7R X5R, X7R
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LT3474
APPLICATIO S I FOR ATIO
Diode Selection
The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from:
ID( AVG) =
IOUT ( VIN - VOUT ) VIN
The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to one half the typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. If using the PWM mode of the LT3474, select a diode with low reverse leakage.
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Table 4 lists several Schottky diodes and their manufacturers.
Table 4. Schottky Diodes
PART NUMBER On Semiconductor MBR0520L MBR0540 MBRM120E MBRM140 Diodes Inc. B0530W B120 B130 B140 HB 10BQ030 30 20 30 40 30 0.5 1 1 1 1 430 500 500 530 420 20 40 20 40 0.5 0.5 1 1 385 510 620 530 550 VR (V) IAVE (A) VF at 0.5A (mV) VF at 1A (mV)
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International Rectifier
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LT3474
APPLICATIO S I FOR ATIO
BOOST and BIAS Pin Considerations
The capacitor and internal diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a 0.22F capacitor will work well. Figure 3 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for full efficiency. For outputs of 2.8V or higher, the standard circuit (Figure 3a) is best. For lower output voltages, the BIAS pin can be tied to the input (Figure 3b). The circuit in Figure 3a is more efficient because the BOOST pin current comes from a lower voltage source. Finally, the BIAS pin can be tied to another source that is at least 3V (Figure 3c). For example, if a 3.3V source is on whenever the LED is on, the BIAS pin can be connected to the 3.3V output. In any case, be sure that the maximum voltage at the BOOST pin is both less than 51V and the voltage difference between the BOOST and SW pins is less than 25V.
VIN
BIAS BOOST LT3474 VIN SW GND
C3 VOUT
3474 F03a
VBOOST - VSW VOUT MAX VBOOST VIN + VOUT
(3a)
VIN
BIAS BOOST LT3474 VIN SW GND
C3 VOUT
3474 F03b
VBOOST - VSW VIN MAX VBOOST 2VIN
(3b)
VIN2 > 3V BIAS BOOST LT3474 VIN SW GND
3474 F03c
C3 VOUT
VIN
VBOOST - VSW VIN2 MAX VBOOST VIN2 + VIN MINIMUM VALUE FOR VIN2 = 3V
(3c)
Figure 3. Generating the Boost Voltage
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Programming LED Current The LED current can be set by adjusting the voltage on the VADJ pin. For a 1A LED current, either tie VADJ to REF or to a 1.25V source. For lower output currents, program the VADJ using the following formula:
ILED = 1A * VADJ 1.25V
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Voltages less than 1.25V can be generated with a voltage divider from the REF pin, as shown in Figure 4.
REF R1 VADJ R2 GND
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LT3474
Figure 4. Setting VADJ with a Resistor Divider
In order to have accurate LED current, precision resistors are preferred (1% or better is recommended). Note that the VADJ pin sources a small amount of bias current, so use the following formula to choose resistors:
R2 =
VADJ 1.25V - VADJ + 50nA R1
To minimize the error from variations in VADJ pin current, use resistors with a parallel resistance of less than 4k. Use resistors with a series resistance of 5.11k or greater so as not to exceed the 250A current limit on the REF pin.
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LT3474
APPLICATIO S I FOR ATIO
Dimming Control There are several different types of dimming control circuits. One dimming control circuit (Figure 5) changes the voltage on the VADJ pin by tying a low on-resistance FET to the resistor divider string. This allows the selection of two different LED currents. For reliable operation, program an LED current of no less than 35mA. The maximum current dimming ratio (IRATIO) can be calculated from the maximum LED current (IMAX) and the minimum LED current (IMIN) as follows: IMAX = IRATIO IMIN Another dimming control circuit (Figure 6) uses the PWM pin and an external NFET tied to the cathode of the LED. When the PWM signal goes low, the NFET turns off, turning off the LED and leaving the output capacitor charged. The PWM pin is pulled low as well, which disconnects the VC pin, storing the voltage in the capacitor tied there. When the PWM pin goes high again, the LED
REF R1 VADJ R2 DIM GND
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LT3474
LED
Figure 5. Dimming with an NFET and Resistor Divider
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current returns rapidly to its previous on state since the compensation and output capacitors are at the correct voltage. This fast settling time allows The LT3474 to maintain diode current regulation with PWM pulse widths as short as 40s. The maximmum PWM dimming ratio (PWMRATIO) can be calculated from the maximum PWM period (tMAX) and minimum PWM pulse width (tMIN) as follows: tMAX = PWMRATIO tMIN Total dimming ratio (DIMRATIO) is the product of the PWM dimming ratio and the current dimming ratio. Example: IMAX = 1A, IMIN = 0.1A, tMAX = 12ms, tMIN = 40s
IRATIO = 1A = 10:1 0.1A 12ms = 300:1 PWMRATIO = 40s DIMRATIO = 10 * 300 = 3000:1
PWM 60Hz TO 10kHz
PWM LT3474 GND
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Figure 6. Dimming Using PWM Signal
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LT3474
APPLICATIO S I FOR ATIO
Layout Hints As with all switching regulators, careful attention must be paid to the PCB layout and component placement. To maximize efficiency, switch rise and fall times are made as short as possible. To prevent electromagnetic interference (EMI) problems, proper layout of the high frequency switching path is essential. The voltage signal of the SW and BOOST pins have sharp rise and fall edges. Minimize the area of all traces connected to the BOOST and SW pins and always use a ground plane under the switching regulator to minimize interplane coupling. In addition, the ground connection for frequency setting resistor RT (refer to Figure 1) should be tied directly to the GND pin and not shared with any other component, ensuring a clean, noisefree connection.
VIA TO LOCAL GND PLANE VIA TO OUT
Figure 7. Recommened Component Placement
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PWM SHDN VIN GND
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17
LT3474
TYPICAL APPLICATIO S
Step-Down 1A LED Driver with PWM Dimming
VIN 6V TO 36V C1 2.2F 50V VIN SHDN LT3474 RT REF R1 80.6k C4 0.1F VADJ VC GND LED1 M1 PWM BIAS OUT PWM LED C2 2.2F 6.3V BOOST SW D1 C3 0.22F L1 6.3V 10H
D1: B140HB C1 TO C3: X5R OR X7R LED1: LUMILEDS LXHL-PW09 M1: Si2302ADS
Step-Down 1A LED Driver with Two Series Connected LED Output
VIN 12V TO 36V C3 0.22F L1 10H 6.3V D1 BIAS OUT C2 2.2F 10V LED1 LED2
SHDN LT3474 RT REF R1 33.2k C4 0.1F VADJ VC GND
SW
EFFICIENCY (%)
C1 2.2F 50V
VIN
BOOST
LED
D1: MBRM 140 C1 TO C3: X5R OR X7R LED1, LED2: LUMILEDS LXHL-PW09
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LED Current in PWM Mode
ILED1 500mA/DIV
V(PWM) 5V/DIV
1ms/DIV
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Efficiency, Two LED Output
95 90 85 80 75 70 65 60 55 0 200 400 800 600 LED CURRENT (mA) 1000
3474 G01
VIN = 12V VIN = 24V
3474 TA02
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PACKAGE DESCRIPTIO
2.74 (.108)
6.60 0.10 4.50 0.10
SEE NOTE 4
0.65 BSC
RECOMMENDED SOLDER PAD LAYOUT
4.30 - 4.50* (.169 - .177)
0.09 - 0.20 (.0035 - .0079)
0.50 - 0.75 (.020 - .030)
NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
FE Package 16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation BA
4.90 - 5.10* (.193 - .201) 2.74 (.108) 16 1514 13 12 1110 9
2.74 (.108) 0.45 0.05 1.05 0.10 2.74 6.40 (.108) (.252) BSC
12345678 1.10 (.0433) MAX
0 - 8
0.25 REF
0.65 (.0256) BSC
0.195 - 0.30 (.0077 - .0118) TYP
0.05 - 0.15 (.002 - .006)
FE16 (BA) TSSOP 0204
4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE
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TYPICAL APPLICATIO U
Step-Down 1A LED Driver
VIN 6V TO 36V C3 0.22F L1 10H 6.3V D1 BIAS OUT C2 2.2F 6.3V LED1
C1 2.2F 50V
VIN
SHDN
BOOST SW
LT3474
RT REF
R1 80.6k
VADJ VC GND
C4 0.1F
LED
D1: MBRM 140 C1 TO C3: X5R OR X7R LED1: LUMILEDS LXHL-LW3C
3474 TA02a
RELATED PARTS
PART NUMBER LT1618 LT1766 LT1956 LT1961 LT1976/LT1977 LT3430/LT3431 LT3433 LT3434/LT3435 LTC3453 LT3467/LT3467A LT3477 LT3479 DESCRIPTION Constant Current, 1.4MHz, 1.5A Boost Converter 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 500kHz, High Efficiency Step-Down DC/DC Converter 1.5A (ISW), 1.25MHz, High Efficiency Step-Up DC/DC Converter 60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters with BurstMode(R) Operation 60V, 2.5A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converters 60V, 400mA (IOUT), 200kHz, High Efficiency Step-Up/Step-Down DC/DC Converters with Burst Mode Operation 60V, 2.5A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters with Burst Mode Operation 1MHz, 800mA Synchronous Buck-Boost High Power LED Driver 1.1A (ISW), 1.3MHz/2.1MHz, High Efficiency Step-Up DC/DC Converters with Integrated Soft-Start 3A, 42V, 3MHz Step-Up Regulator with Dual Rail to Rail Current Sense 3A, Full Featured DC/DC Converter with Soft-Start and Inrush Current Protection COMMENTS VIN: 1.6V to 18V, VOUT(MAX) = 36V, IQ = 1.8mA, ISD = <1A, MS10 Package VIN: 5.5V to 60V, VOUT(MAX) = 1.20V, IQ = 2.5mA, ISD = 25A, TSSOP16/E Packages VIN: 5.5V to 60V, VOUT(MAX) = 1.20V, IQ = 2.5mA, ISD = 25A, TSSOP16/E Packages VIN: 3V to 25V, VOUT(MAX) = 35V, IQ = 0.9mA, ISD = 6A, MS8E Package VIN: 3.3V to 60V, VOUT(MAX) = 1.20V, IQ = 100A, ISD = <1A, TSSOP16E Package VIN: 5.5V to 60V, VOUT(MAX) = 1.20V, IQ = 2.5A, ISD = <25A, TSSOP16/E Packages VIN: 4V to 60V, VOUT: 3.3V to 20V, IQ = 100A, ISD = <1A, TSSOP16E Package VIN: 3.3V to 60V, VOUT(MAX) = 1.20V, IQ = 100A, ISD = <1A, TSSOP16E Package VIN: 2.7V to 5.5V, VOUT(MAX) = 5.5V, IQ = 2.5mA, ISD = <6A, QFN Package VIN: 2.4V to 16V, VOUT(MAX) = 40V, IQ = 1.2mA, ISD = <1A, ThinSOTTM Package VIN: 2.5V to 2.5V, VOUT(MAX) = 40V, IQ = 5mA, ISD = <1A, QFN, TSSOP16E Packages VIN: 2.5V to 24V, VOUT(MAX) = 40V, IQ = 6.5mA, ISD = <1A, DFN and TSSOP Packages
Burst Mode is a registered trademark of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation.
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Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
LT/TP 1104 1K * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2004

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